• 专利附录
  • 专利说明
  • 权利要求
  • 类似技术
  • 同族专利
  • 引用文献
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  • 优先权
    • 专利摘要:
    A circuit comprising: an audio amplifier (314) adapted to amplify an input signal (SAUDIO) for generating an output signal (SOUT +, SOUT-) suitable for driving a speaker (316); a first circuit (318) adapted to generate a first analog signal (SI) based on a current level absorbed by the speaker (316); a second circuit (320) adapted to generate a second analog signal (Sy) based on a voltage supplied across the speaker (316); a third circuit (322, 312) adapted to: generate a third analog signal (RESIDUE) based on the difference between the first and second analog signals; and change the input signal (SAUDIO) based on the third analog signal.
    公开号:FR3030983A1
    申请号:FR1462870
    申请日:2014-12-19
    公开日:2016-06-24
    发明作者:Christian Fraisse；Angelo Nagari
    申请人:STMicroelectronics Alps SAS；
    IPC主号:
    专利说明:

    [0001] TECHNICAL FIELD This description relates to the field of systems and methods for protecting audio loudspeakers, and in particular a circuit and a method for protecting a loudspeaker. BACKGROUND OF THE INVENTION audio against undesirable oscillations and against overheating. BACKGROUND Mobile devices such as mobile phones, and in particular smartphones, are increasingly equipped with relatively high power audio amplifiers to drive hands-free speakers and provide high quality audio functionality. Although in an implementation it is possible to use an amplifier of class AB as an audio amplifier, the efficiency of such an amplifier does not generally exceed 20-25% in most practical situations, which leads to a current consumption. high, which is undesirable. Class D amplifiers offer an alternative solution that offers significantly better performance than class AB amplifiers.
    [0002] One difficulty is that the micro-speakers that are commonly used in mobile devices are relatively fragile and can easily be damaged. A typical micro-speaker can typically withstand a continuous power of less than half a watt. In fact, micro-speakers generally suffer from two main functional limitations.
    [0003] First, there is a limit in the permissible range of the speaker diaphragm excursion without damaging the diaphragm. For a typical micro-speaker, the maximum allowable excursion is about 0.4 mm. However, the loudspeakers generally have a mechanical resonance frequency of about 1 kHz, and at this frequency the speaker swing limit can be exceeded with a relatively low power signal. Although a high pass filter could be used to attenuate the signal energy at and below this resonance frequency, this would have a negative effect on sound quality. In addition, the mechanical resonance frequency can change significantly under varying operating conditions, for example as a function of temperature, aging, and external forces such as blocking speaker access, which means that a filter 20 should suppress a relatively wide frequency band. Second, overheating can damage the speaker. Overheating occurs when the speaker is supplied with more power than it can dissipate. If, for example, air movement around the loudspeaker is prevented by clogged speaker access, speaker cooling becomes less efficient, and overheating can occur within seconds at a power level. relatively weak. Existing solutions to protect the micro-speakers from damage tend to be inappropriate and / or complex. There is therefore a need in the art for an improved speaker protection system and method. SUMMARY An object of embodiments of the present disclosure is to at least partially meet one or more needs of the prior art. According to one aspect, there is provided a circuit comprising: an audio amplifier adapted to amplify an input signal to generate an appropriate output signal for driving a speaker; a first circuit adapted to generate a first analog signal based on a current level absorbed by the speaker; a second circuit adapted to generate a second analog signal based on a voltage supplied across the loudspeaker; a third circuit adapted to: generate a third analog signal based on the difference between the first and second analog signals; and modify the input signal based on the third analog signal. According to one embodiment, at least one of the first and second analog signals is normalized with respect to the other. According to one embodiment, the second circuit comprises an analog filter adapted to shift the phase of the output signal to generate the second analog signal. According to one embodiment, the input signal is modified by subtracting the third analog signal from the input signal. According to one embodiment, the circuit further comprises a compressor adapted to partially compress the third analog signal before subtracting it from the input signal. According to one embodiment, the compressor is adapted to make the third analog signal null when the first analog signal is greater than the second analog signal. According to one embodiment, the compressor comprises one or more variable resistors in a conduction path of the third analog signal, the resistance of said one or more variable resistors being iteratively selected based on said difference. According to one embodiment, the audio amplifier is a class D audio amplifier. In another aspect there is provided a system comprising: a speaker; a processing device adapted to generate a digital audio stream; a digital-to-analog converter adapted to generate the input signal based on the digital audio stream; and the aforementioned circuit adapted to drive the speaker based on the input signal.
    [0004] In another aspect there is provided a method comprising: amplifying, by an audio amplifier, an input signal to generate an appropriate output signal for driving a speaker; generating, by a first circuit, a first analog signal based on a current level absorbed by the speaker; generating, by a second circuit, a second analog signal based on a voltage supplied across the loudspeaker, at least one of the first and second analog signals being normalized with respect to the other; generating, by a third circuit, a third analog signal based on the difference between the first and second analog signals; and modifying, by the third circuit, the input signal on the basis of the third analog signal. BRIEF DESCRIPTION OF THE DRAWINGS The above-mentioned and other features and advantages will become apparent from the following detailed description of embodiments, given by way of illustration and not limitation, with reference to the accompanying drawings, in which: FIG. 1 schematically illustrates an equivalent electrical circuit of a loudspeaker; Fig. 2A is a graph illustrating an example of the typical relationship between the level of current and the frequency in a loudspeaker; Fig. 2B is a graph illustrating an example of the typical relationship between the phase of the current and the frequency in a loudspeaker; FIG. 3 schematically illustrates an audio system comprising a loudspeaker protection system according to an exemplary embodiment of the present description; FIG. 4 schematically illustrates an amplification and protection system of FIG. 3 in more detail according to an exemplary embodiment; FIG. 5 diagrammatically illustrates the amplification and protection system 5 of FIG. 3 in more detail according to another exemplary embodiment; Figure 6 schematically illustrates a compression and control circuit of Figure 5 in more detail according to an exemplary embodiment; and Fig. 7 is a graph showing the feedback gain based on the control of the compressor of Fig. 6 according to an exemplary embodiment. DETAILED DESCRIPTION In the present description, the term "connected" is used to denote a direct connection between two elements, while the term "coupled" is used to designate a connection between two elements which may be direct, or to be intermediate of one or more other components such as resistors, capacitors or transistors.
    [0005] The publication of A. Nagari et al. titled "An 8 IF 2.5W 1% -THD 10 4 dB (A) -Dynamic-Range Class-D Audio Amplifier With Ultra-Low EMI System and Current Sensing for Speaker Protection", IEEE Journal of Solid-State Circuits, Vol. 47, No. 12, December 2012, discloses a speaker protection circuit that uses an analog-to-digital converter to convert current measurements in a speaker to digital values, and a digital signal processor (DSP) for process the digital values and adjust, in response, the digital signal supplied to the loudspeaker.
    [0006] Although the solution described in this publication is relatively efficient, using a DSP results in relatively high power consumption, uses a relatively large area, and adds complexity. In addition, the concatenation of the DSP software to protect the speaker with manufacturer-specific software to perform other DSP functions is a complex task. FIG. 1 schematically illustrates an equivalent electrical circuit 100 of a loudspeaker, which is for example a micro-speaker of a mobile telephone, having two input terminals 102 and 104 receiving signals Sour + and SouT_ of an amplifier (not shown in Figure 1). The circuit 100 comprises, coupled to the terminal 102 of the loudspeaker, a resistor 106 representing an equivalent resistive load that varies with the temperature. The resistor 106 and coupled in series with an inductor 108. The loudspeaker also comprises a resonant part modeled by the parallel connection of a capacitor 110, a resistor 112 and an inductor 114, each of them being coupled between the inductor 108 and the terminal 104.
    [0007] The resonant part represents the resonant behavior of the speaker around a certain frequency, which varies with temperature and other effects. Assuming that the loudspeaker has a nominal impedance of about 8 ohms, the resistor 106 has for example a resistance of about 7.4 ohms, the inductance 108 has an inductance of between 40 and 100 pH, for example At about 60 pH, the capacitor 110 has a capacity of about 160 pF, the resistor 112 has a resistance of about 16 ohms, and the inductor 114 has an inductance of about 200 pH. As used herein, the term "about" is used to refer to a range of +/- 10 percent. Of course, the aforementioned values are only examples, and such a model will vary considerably from one speaker to another, depending on characteristics such as speaker size and impedance.
    [0008] FIG. 2A illustrates an example of the relationship between the current level (CURRENT LEVEL), in decibels, and the frequency, on a logarithmic scale in kHz, in the loudspeaker represented by the model of FIG. illustrated in FIG. 2A, the current level remains, for example, constant at 0 dB for low frequencies up to the resonance frequency fR of the loudspeaker, where the level of the current drops sharply . In the example of Figure 2A, the resonant frequency fR is just below 1 kHz, and at this point the current reaches for example a low point of -10 dB. Above the resonant frequency, the current level rises to the 0 dB level before falling again as the frequency approaches the speaker limit at about 20 kHz. According to the embodiments described herein, to avoid damaging the loudspeaker, the signal energy at about the resonant frequency fR is, for example, attenuated. FIG. 2B illustrates an example of the relationship between the phase of the current (CURRENT PHASE), in degrees, and the frequency, on a logarithmic scale in kHz, in the loudspeaker represented by the model of FIG. In FIG. 2B, a dotted line illustrates a case in which the loudspeaker does not suffer from having a resonant frequency, and in such a case the phase, for example, shifts progressively with the frequency, reaching for example a phase shift of -30. degrees at 20 kHz. However, as shown by a solid line curve in FIG. 2B, in the case where the loudspeaker has a resonant frequency fR, the phase shift for example decreases rapidly near this frequency. Just below the resonance frequency fR, the phase shift for example reaches a peak of about -30 degrees, then suddenly switches to a positive phase shift with a peak of 30 degrees just above the resonance frequency fR. The phase shift then decreases progressively as the frequency increases, becomes negative, and reaches the -30 degree level at 20 kHz. Figure 3 schematically illustrates an audio system 300 according to an exemplary embodiment of the present description. For example, an audio digital audio signal is provided on a connection 302 from a digital signal processor (DSP) 304, for example on the basis of an audio stream 306 (AUDIO) from an audio file stored in an audio file. memory (not shown in Figure 3). The connection 302 is coupled to an audio amplifier block (AUDIO AMP) 308, and in particular to a digital-to-analog converter (DAC) 310, which converts the AUDIO signal into an analog voltage signal SAUDIO. In some embodiments, an analog signal S AUDIO 'may instead or additionally be provided via an input 311 of the audio amplifier block 308 from an analog source. The analog signal S AUDIO is supplied to a power limiting element 312, which adjusts the power of the signal supplied to the loudspeaker based on a feedback path described hereinafter. The feedback signal, called RESIDUE in FIG. 3, corresponds, for example, to the time convolution of the speaker current IspKR and the voltage VspKR, with appropriate weighting and appropriate phase adjustment. In particular, the time convolution VspKR-I SPKR corresponds to the impedance ZspKR in the frequency domain. For example, the feedback signal RESIDUE is subtracted from the signal S AUDIO by the element 312. This subtraction of the impedance ZspKR from the signal path leads to a real-time equalization of the path, itself weighted by the response frequency of the impedance of the speaker. The output of the power limiting element 312 is supplied to an audio amplifier 314, which is for example a class D amplifier, although in alternative embodiments it is possible to use other types of amplifiers, such as a power amplifier. class AB. The amplifier 314 supplies the differential output signals SuT +, SouT_ to the loudspeaker 316, which is for example a micro-loudspeaker. A current measurement is made on one of the output lines of the audio amplifier 314, and is supplied to a circuit 318 which generates an analog current signal SI, represented for example by a voltage signal, based on the current level in the speaker. In addition, a voltage measurement is made between the output lines of the audio amplifier 314, and is supplied to a circuit 320, which provides an analog signal Sv based on the difference of voltage VOUT between SouT + and SouT_. The current measurement is, for example, independent of the DC resistance of the loudspeaker, in other words it is normalized to assume a unit resistance. For example, in one embodiment, the current measurement is based on the voltage drop in a resistor in one of the output lines of the audio amplifier 314. This resistance has, for example, a resistor R SENSE equal to one. G SENSE fraction of the DC resistance of the loudspeaker. Thus in DC conditions, the current measurement will be equal to GSENSE * VOUT. In addition, at least one of the current and voltage signals SI, Sv is for example normalized by the circuit 318 or 320, to cancel the gain G SENSE applied to the current measurement, so that under current conditions Continuous current and voltage signals will be substantially equal. A difference between the current and voltage signals SI, Sv is then generated for example by a circuit 322, for example by subtracting the current signal from the voltage signal. The result constitutes the RESIDUE signal to subtract S AUDIO input signals by the element 312. In some embodiments, the RESIDUE signal may be generated by applying, by the circuit 322, a gain to the difference between the signals. current and voltage. Furthermore, in some embodiments, the RESIDUE signal is activated only in the case where the voltage signal Sv is greater than the current signal SI, and in the other case, the signal RESIDUE and zero. In operation, the analog feedback loop formed by the circuits 318, 320 and 322 and the element 312 provides an almost instantaneous correction of the signal energy level if an impedance increase of the loudspeaker 316 is detected. in other words if the voltage signal Sv becomes greater than the current signal S.
    [0009] FIG. 4 schematically illustrates the audio amplifier block 308 of FIG. 4 in more detail according to an exemplary embodiment. As shown in FIG. 4, the analog audio signal S AUDIO of FIG. 3 is for example a differential signal S AUDIO +, SAUDIO- supplied to the corresponding input nodes 403 and 404 of the audio amplifier 314. The signal S AUDIO can be provided by differential outputs of the DAC 310 (not shown in Figure 4), or through input lines 10 forming the input 311 of Figure 3 (also not shown in Figure 4). In the example of FIG. 4, the audio amplifier 314 is a class D audio amplifier comprising an integrator 405. A positive output 406 of the integrator 405 is coupled to an input of a comparator 408, whose Another input is coupled to receive a double-ramp signal (D-RAMP). Similarly, a negative output 407 of the integrator 405 is coupled to an input of a comparator 409, the other input of which is coupled to receive the double ramp signal. The output of each comparator 408, 409 is coupled to an H bridge 410. The H bridge 410 includes a conductor 412 coupled to an output 413 of the audio amplifier block 308, and a conductor 414 coupled to an output 415 of the block Audio amplifier 308. The leads 412 and 414 are coupled to the supply voltage VDD through respective switches. The conductor 412 is also coupled to a node 416 via another controller, and the conductor 414 is also coupled to a node 418 via yet another switch. Node 416 is coupled to ground via resistor 420, and node 418 is coupled to ground through resistor 422. Each of resistors 420, 422 has a resistor Rsense. The outputs of the comparators 408, 409 control the switches of the H-bridge so that the output 413 is coupled to the supply voltage VDD and the output 415 is coupled to the ground by intermediate of the resistor 422, or such that the output 413 is coupled to ground through the resistor 420 and the output 415 is coupled to the supply voltage VDD.
    [0010] The circuit 318 has inputs coupled to the nodes 416 and 418. In one embodiment, the gain GI of an amplifier stage of the circuit 318 is equal to: GI = Gtrim GRdc where Gtrim is a value, for example selected from a table of correspondence, intended to compensate for differences between each of the resistors 420 and 422 and their nominal value, and GRdc is a value, for example selected in another correspondence table, used to compensate for differences between the actual DC resistance of the top 316 and its nominal value. The circuit 318 also comprises for example a first order filter for filtering the analog current signal. A first order filter is for example sufficient since the current is also filtered by the load. The circuit 320 for generating the analog voltage signal Sv comprises, for example, an amplifier stage 320A having, for example, a GV gain of approximately 0.125. The circuit 320A also includes for example a second order filter for filtering the analog voltage signal. The circuit 320 also comprises, for example, a variable RO filter 320B consisting of a variable resistor 424 coupled between an output of the differential amplifier stage 320A and a node 426 of a variable resistor 428 coupled between the other output of the amplifier. differential amplifier stage 320A and a node 430, and a capacitor 432 coupled between the nodes 426 and 430. The resistors of the variable resistors 424, 428 are for example selected such that the RC 320B filter compensates for the phase difference between the voltage between the output terminals 413, 415, and the current in the speaker 316, which may vary depending on the inductance of the loudspeaker 316. For example, if B13461 - 14-GR2-0088 12 is calls Rle the resistance of each of the resistors 424, 428, the value of Rle is for example adapted to be equal to: Rle = Le / (40x10-12 * Rdc), where Le is the inductance of the loudspeaker 316 like this is represented by inductance 108 in FIG. 1, and Rdc is the DC resistance of the loudspeaker 316. Taking the example in which Le is equal to 60 pH and Rdc is equal to 7.4 ohms, the resistance Rle is for example chosen to be equal to 202, 7 kilo-ohms. In an environment, each of the resistors 424, 428, for example, is variable around a value of about 203 kilo-ohms, and the capacitor 465, for example, has a capacitance of about 20 pF. FIG. 5 diagrammatically illustrates the audio amplifier block 308 of FIG. 3 in greater detail according to an alternative embodiment with respect to FIG. 4. Many elements of the circuit of FIG. 5 are identical to those of the circuit of FIG. the same numerical references and will not be described again in detail. FIG. 5 illustrates the integrator 405 in an example 20 where it comprises a differential amplifier 501 having its positive input coupled to an output of the power limiter element 312 through a resistor 502 and its negative input coupled to the other output of the power limiting element 312 through a resistor 504. The differential amplifier 501 also has its positive input coupled to its negative output by a capacitor 506, and at the output node 413 of the Audio amplifier block 308 via a resistor 508. Similarly, the differential amplifier has, for example, its negative input coupled to its positive output by a capacitor 510, and to the output node 415 of the audio amplifier block 308 by an amplifier. 512. In addition, in the embodiment of FIG. 5, the output lines of the circuit 322 are coupled to the power limiting element 312 via of a compressor 514, which is for example controlled by a control block (CTRL) B13461 - 14 - GR2-0088 13 516. The control block 516 for example selects a resistance value of the compressor 514 on the basis of the signal of residual, the analog current signal SI supplied to the output of the circuit 318, and the analog voltage signal Sv supplied to the output of the circuit 320. FIG. 6 schematically illustrates the compressor 514 and the control block 516 in more detail according to an exemplary embodiment. The compressor 514 comprises for example a selection circuit 601, and a variable resistor consisting of a series 602 of 63 resistors coupled to a node 603, and another variable resistor consisting of a series 604 of 63 resistors coupled to a node 605. The node 603 is for example coupled to an output 606 of the compressor 514 via a resistor 607, and the node 605 is for example coupled to an output 608 of the compressor 514 via a resistor 609. Although in some embodiments the resistors of each of the series 602, 604 each have the same resistance, in the example of FIG. 6, the resistance of each series closest to the node 603, 604 has, for example, a relatively high resistance. low, and the resistance of the other elements gradually increases as they move away from the nodes 603, 605. A node 610 at the inlet of the compressor 514, which is coupled to an output of the circuit 322, may be selectively coupled by one of a series of 64 switches to any node of the series of resistors 602. These switches are respectively controlled by 1-bit control signals COMPO1 to COMP64. Similarly, a node 611 at the input of the compressor 514, which is coupled to the other output of the circuit 322, can be selectively coupled by one of a series of 64 switches to any node in the series of resistors 604. These switches are also respectively controlled by the control signals COMPO1 to COMP64.
    [0011] B13461 - 14-GR2-0088 14 The selection block 601 activates one or both of the control signals COMPO1 to COMP64 based on a selection signal S supplied by the control circuit 516. An example of the feedback gain provided by the selection of each of the control signals is illustrated in FIG. 7. FIG. 7 is a graph showing the feedback gain (FEEDBACK GAIN) in decibels as a function of the control signal (FIG. CONTROL) applied to the compressor. In the case where none of the COMPO1 to COMP64 'comuno signals is activated, the loop is for example open. The lowest selectable gain corresponding to the activation of the signal COMPO1 is for example about -32 dB, while the highest gain corresponding to the activation of the control signal COMP64 is for example about +6 dB. The gain decreases, for example, relatively linearly for the control signals COMP64 to COMP20, then decreases exponentially to the minimum value. How resistance values in the compressor and other resistors in the circuit can be selected to achieve the feedback gains shown in Figure 7 will be readily apparent to those skilled in the art. In addition, it will be apparent to those skilled in the art that a similar compression law, having a curve similar to that of FIG. 7, could be obtained with a larger or smaller number of control signals. Referring again to FIG. 6, the control circuit 516 comprises, for example, a conversion stage 612, which receives the RESIDUE residue signal from the input nodes 610, 611, and provides a differential mode conversion to the Non-differential mode. The stage 612 comprises for example a filter RC consisting of a resistor 614 coupling the line 610 to a node 615, a resistor 616 coupling the line 611 to a node 617, and a capacitor 618 coupled between the nodes 615 and 617. Resistors 614 and 616, for example, have resistances of about 400 kilo-ohms, and capacitor 618, for example, has a capacitance of about 15 pF. The node 615 is coupled to a negative input of a differential amplifier 620 via a resistor 622 having for example a resistance of about 400 kilo-ohms. The differential amplifier 620 for example has a feedback path between its output and its negative input comprising a resistance of about 800 kilo-ohms. Similarly, the node 617 is for example coupled to a negative input of a differential amplifier 624 via a resistor 626 having a resistance of about 400 kilo-ohms.
    [0012] The differential amplifier 624 for example has a feedback path between its output and its negative input comprising a resistance of about 800 kilo-ohms. The positive inputs of the differential amplifiers 620, 624 are coupled to a reference voltage level equal, for example, to about half of a common mode voltage VCM of the differential signals. Each of the outputs of the differential amplifiers 620, 624 is for example coupled, via a respective resistor 627, 628, to a node 630. The node 630 is for example coupled to a positive input of a differential amplifier 632 having its output coupled to ground by the series connection of two resistors 633, 634 having for example resistances of about 1200 and 400 kilo-ohms respectively. An intermediate node between the resistors 633 and 634 is coupled to the negative input of the differential amplifier 632. The amplifier 632 provides the absolute value, for example the RMS (RMS value), of the differential input residual. 612. The output of the block 612 is coupled to the positive input of a comparator 636, which compares the signal to a reference voltage VREF generated by the series connection of a resistor 637 and a variable resistor. 638 coupled between the supply voltage VDD and the ground. In one embodiment, the supply voltage VDD is about 2 V, the resistor 637 has a resistance of about 63 kilo-ohms, and the resistor 638 has a resistance of about 27 kilo-ohms. so that the reference VREF B13461 - 14-GR2-0088 16 is about 0.54 V. The comparator 636 provides on its output a binary value, which is for example high if the signal from the block 612 is greater than VREF, and low if the signal from block 612 is less than VREF.
    [0013] The output of the comparator 636 is coupled to an accumulator 640, which is for example 12 bits. The accumulator 640 comprises for example a 12-bit adder 641, which receives the output binary signal from the comparator 636, and adds it to the signal S present at the output of the accumulator 640 to generate a modified value S ' The flip-flop 642 is clocked by a clock signal CLK, which for example has a frequency of about 48 kHz. The flip-flop 642 provides, for example, the selection signal S on its output. In one embodiment, the selection circuit 601 generates the control signals CO1'4P01 through COMP64 based on the six most significant bits (MSB) of the S value of 12 bits. For example, when the MSBs are all zero, none of the COMPO1 to COMP64 signals are turned on, whereas when the six MSBs are all one, the COMP64 signal is selected. The adder 641 is for example adapted to add 32 to the value of 12 bits, in other words the binary word "0000 0010 0000", if the output of the comparator 636 is high, and to subtract 1 from the value of 12 bits, in other words subtract the binary word "0000 0000 0001" if the output of the comparator 636 is low.
    [0014] Flip-flop 642 is for example reset to zero by a reset signal R provided by a sign verification block which will now be described. This block activates for example the reset signal R if the current signal SI is greater than the voltage signal Sv, so that the signal on the output of the compression block 508 is brought to the low state. The differential current signal SI is for example supplied to a differential mode conversion block in non-differential mode 643, which is for example similar to block 612, and which will not be described again a detail. Similarly, the differential voltage signal Sv is, for example, provided by a differential value conversion block in absolute value (for example RMS) 644, which is also for example similar to block 612. , and that will not be described again in detail. The outputs of the blocks 643 and 644 are supplied by respective resistors 646 and 648 respectively to positive and negative inputs of a comparator 650. The positive and negative inputs are also, for example, respectively coupled to ground via capacitors 652 and 654 respectively. Resistors 646, 648 each have, for example, a resistance of about 64000 kilo-ohms, and capacitors 652, 654 each have, for example, a capacitance of about 40 pF. The comparator 650 compares for example the input signals, and activates the reset signal R if the current signal is greater than the voltage signal.
    [0015] An advantage of the compression system depicted in FIG. 5 is that it will tend to reduce the signal-to-noise ratio in the system. This allows, for example, the amplifier 314 to be relieved in terms of noise reported at the input, thereby reducing power consumption and the chip surface. The 6-bit control provided by the compression and control circuits 514, 516 limits the SNDR (signal-to-noise ratio plus distortion) to 36 dB around the resonant frequency, which corresponds to a dynamic range of 46 dB, which 25 which is for example sufficient to ensure a significant effect of psycho-acoustic masking of sound pressure level. In alternative embodiments, compression based on more or less bits may be applied. For example, a 7-bit compressor will give a 46 dB SNDR, and a dynamic range of 52 dB. An advantage of the embodiments described herein is that the circuit provides a simple and effective control mechanism for protecting a loudspeaker from damage associated with overheating and / or displacement beyond its maximum allowable excursion limits. The inventors have found that the implementation of such a circuit adds relatively little current consumption, and a limited area for the additional circuit. Although certain specific embodiments of the invention have been described and shown in the figures, it will be apparent to those skilled in the art that many modifications and alterations could be applied. For example, although the figures include many examples of resistances, capacitances and inductances, it will be apparent to those skilled in the art that many different values can be used depending on the specific application. . In addition, it will be apparent to those skilled in the art that the embodiments described herein could be adapted to non-differential instead of differential implementation.
    权利要求:
    Claims (10)
    [0001]
    REVENDICATIONS1. A circuit comprising: an audio amplifier (314) adapted to amplify an input signal (S AUDIO) for generating an output signal (SuT +, SOUT-) suitable for driving a loudspeaker (316); a first circuit (318) adapted to generate a first analog signal (SI) based on a current level absorbed by the speaker (316); a second circuit (320) adapted to generate a second analog signal (Sv) based on a voltage supplied to the terminals of the speaker (316); a third circuit (322, 312) adapted to: - generate a third analog signal (RESIDUE) on the basis of the difference between the first and second analog signals; and 15 - modify the UDIO input signal based on the third analog signal.
    [0002]
    2. Circuit according to claim 1, wherein at least one of the first and second analog signals (SI, Sv) is normalized with respect to the other. 20
    [0003]
    3. Circuit according to claims 1 or 2, wherein the second circuit (320) comprises an analog filter (320B) adapted to shift the phase of the output signal to generate the second analog signal (Sv).
    [0004]
    The circuit of any one of claims 1 to 3, wherein the input signal (S AUDIO) is changed by subtracting the third analog signal (RESIDUE) from the input signal.
    [0005]
    The circuit of claim 4, further comprising a compressor (514) adapted to partially compress the third analog signal (RESIDUE) before subtracting it from the input signal (S AUDIO).
    [0006]
    The circuit of claim 5, wherein the compressor (514) is adapted to nullify the third analog signal (RESIDUE) when the first analog signal is greater than the second analog signal.
    [0007]
    7. Circuit according to claims 5 or 6, wherein the compressor (514) comprises one or more variable resistors (602, 604) in a conduction path of the third analog signal, the resistance of said one or more variable resistors being selected so iterative on the basis of said difference.
    [0008]
    The circuit of any one of claims 1 to 7, wherein the audio amplifier (314) is a class D audio amplifier.
    [0009]
    9. System comprising: a speaker (316); a processing device (304) adapted to generate a digital audio stream (AUDIO); a digital to analog converter (310) adapted to generate the input signal (S AUDIO) based on the digital audio stream; and the circuit of any one of claims 1 to 8 adapted to drive the speaker (316) based on the input signal (S AUDIO) -
    [0010]
    A method comprising: amplifying, by an audio amplifier (314), an input signal (S AUDIO) to generate an output signal (SuT +, SouT_) suitable for driving a loudspeaker (316); generating, by a first circuit (318), a first analog signal (SI) based on a current level absorbed by the speaker (316); generating, by a second circuit (320), a second analog signal (Sv) based on a voltage supplied across the loudspeaker (316), at least one of the first and second analog signals being normalized by relating to the other; generating, by a third circuit (322, 312), a third analog signal (RESIDUE) on the basis of the difference between the first and second analog signals; and modifying, by the third circuit (322, 312), the input signal (S AUDIO) on the basis of the third analog signal (RESIDUE).
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    同族专利:
    公开号 | 公开日
    US9693138B2|2017-06-27|
    CN105721982A|2016-06-29|
    FR3030983B1|2018-02-16|
    EP3035704B1|2019-01-02|
    US20170257701A1|2017-09-07|
    CN105721982B|2019-11-19|
    EP3035704A1|2016-06-22|
    US20160182999A1|2016-06-23|
    CN205430581U|2016-08-03|
    US10021482B2|2018-07-10|
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    法律状态:
    2015-11-23| PLFP| Fee payment|Year of fee payment: 2 |
    2016-06-24| PLSC| Search report ready|Effective date: 20160624 |
    2016-11-21| PLFP| Fee payment|Year of fee payment: 3 |
    2017-11-21| PLFP| Fee payment|Year of fee payment: 4 |
    2019-11-20| PLFP| Fee payment|Year of fee payment: 6 |
    2021-09-10| ST| Notification of lapse|Effective date: 20210806 |
    优先权:
    申请号 | 申请日 | 专利标题
    FR1462870|2014-12-19|
    FR1462870A|FR3030983B1|2014-12-19|2014-12-19|SYSTEM AND METHOD FOR AUDIO SPEAKER PROTECTION|FR1462870A| FR3030983B1|2014-12-19|2014-12-19|SYSTEM AND METHOD FOR AUDIO SPEAKER PROTECTION|
    EP15182698.9A| EP3035704B1|2014-12-19|2015-08-27|System and method for protecting an audio speaker|
    US14/838,437| US9693138B2|2014-12-19|2015-08-28|Audio speaker protection system and method|
    CN201510587716.0A| CN105721982B|2014-12-19|2015-09-15|The system and method for audio tweeter protection|
    CN201520715282.3U| CN205430581U|2014-12-19|2015-09-15|Circuit and system of audio tweeter protection|
    US15/598,100| US10021482B2|2014-12-19|2017-05-17|Audio speaker protection system and method|
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